PHILIPS SOZ 294

GENERAL DESCRIPTION


TYPE SOZ 294/00, HIGH-EFFICIENCY, 100 KW SHORT WAVE BROADCAST TRANSMITTER

 [by: H.E.Eckhardt and A.G.Robeer] - [source: Philips Telecommunication Review Vol.19, No.3, April 1958]

photo Philips SOZ 294 Summary

A 100 kW short wave broadcast transmitter covering the frequency range from 5.9 to 26.1 Mc/s. Modulation is in the anode circuit of the RF power amplifier. Application of high mutual-conductance tubes has resulted in a reduction of the number of stages of RF power amplification to three and of the number of stages of AF amplification to four. Including the high-voltage rectifier for the power amplifiers, only four rectifiers are needed to supply power to all stages of the transmitter. Cooling arrangements have been simplified by the use of air-cooled tubes throughout. Compared with previous designs, floorspace and power consumption requirements have been considerably reduced. The final RF amplifier is of the grounded-cathode type. Operation of the amplifier is stable throughout the frequency range of the transmitter for a single setting of the neutralizing capacitors. Owing to the high efficiency of the RF power stage the overall efficiency is high.

Introduction

In comparing the design of the 100 kW short wave broadcast transmitter treated in the present paper with that of the 100 kW transmitter described in an earlier article, it will be seen that a number of principles embodied in the latter have been retained, but that some notable improvements have been introduced.

In the first place, modern tetrodes have become available with a high mutual conductance, needing very small driving power and, as a result, it has been possible to reduce the preliminary stages to about half their original number. Floor space requirements have consequently been reduced considerably and the number of types of transmitting tubes has become much smaller. Further, the number of automatic tuning mechanisms needed for automatic frequency changing has been brought back to less than half.

In the second place, the use of air-cooled transmitting tubes in all stages of the transmitter has not only greatly simplified the cooling system, but has also resulted in an improvement of the over-all efficiency of the equipment. Some special problems which the transition from water to air cooling created in relation to the anode dissipation of the modulator tubes at the higher modulation frequencies and to the cooling of the anode circuit of the high-frequency power stage, will be discussed when the various transmitter stages are described.

Special cooling of the glass-to-metal seals of the grids and anodes of the RF power tubes has reduced the temperature of these seals considerably and has eliminated the necessity for derating the anode voltage of the power tubes at the higher carrier frequencies. As a result it has been possible to maintain the output power of the transmitter at 100 kW throughout the range from 5.9 to 26.1 Mc/s.

Concurrently with the changes indicated above, the front panel layout of the transmitter has been entirely changed and considerably simplified. Accessibility of the various transmitter components has been a great deal improved.

Philips SOZ 294 fig. 1 Philips SOZ 294 fig. 2 Philips SOZ 294 fig. 3 General arrangement of the transmitter

Both electrically and mechanically the arrangement of the transmitter is characterized by great simplicity. Like its predecessor the new transmitter is arranged for the automatic selection of six spot frequencies within the band from 5.9 to 26.1 Mc/s. Fig. 1 shows a block diagram of the various units composing the transmitter, while Figs.2 and 3 respectively show a groundplan layout and a front view of one of the transmitters as installed at the World Broadcasting Centre of the Netherlands PTT Administration.

In a centre of this nature it is convenient to concentrate the frequency generating equipment for all transmitters at one point where favourable conditions for stable and continuous operation of the oscillators can be easily maintained. As shown on Fig. 1, the oscillators determining the various carrier frequencies do not, therefore, form part of the transmitter proper. Eight of these oscillators are provided, viz six regulars and two spares and the cabinet housing the two preliminary power amplification stages of the transmitter accordingly contains eight pre-tuned units, each comprising a buffer and a frequency doubler stage. Selection of the proper unit is effected by means of an eight-point switch which, like the tuning elements of the transmitter, is driven by an automatic Instantuner.

Amplification of the 6 watt output signal of the frequency doubler, up to the carrier output power of 100 kW, takes place in only three stages. For the two preliminary power stages the necessary amount of amplification has been attained by the use of parallel-connected tetrodes. Power stage I contains four double tetrodes, type QQE 06/40, connected in push-pull-parallel, and drives the second power stage with 4 tetrodes, type QBL 5/3500, also in push-pull-parallel. This latter stage can deliver an output of 10 kW, which is considerably more than required by the final power stage for the full output power of 100 kW.

The final power amplifier is equipped with two type TBL 12/100 tubes. These are air-cooled triodes with thoriated tungsten filaments and a maximum anode dissipation of 45 kW. With thoriated filaments emissions of 40 to 100 mA per Watt cathode heating power can be reached, which is considerably more than the 4 to 5 mA per Watt for pure tungsten filaments, and permits the cathode heating power to be greatly reduced. For a filament current of 196 A at 17.5 V the peak emission current allowed during operation is 120 A, the saturation current being over 250 A.

As an example, a tube setting for 26 Mc/s is given below:

carrier frequency ƒ  =  26  Mc/s
anode voltage Va  =  10.4  kV
grid voltage Vg  =  -1050  V
anode current Ia  =  2 x 6.1  A
grid current Ig  =  2 x 1.9  A
anode dissipation Wia  =  2 x 12  kW
output on dummy load Wo  =  103  kW
efficiency, covering losses in anode
circuit and harmonic suppressor
η
 =
 81
 %

Owing to the high mutual conductance of these tubes, a large power amplification factor can be attained in a single stage. Connection between the amplifier output circuit and the aerial feeder is established via a harmonic suppression filter, comprising a single balanced π-section, and a television interference filter. Behind these filters, the output can be connected either to the aerial feeder or to the artificial aerial for testing purposes. The power amplifier, filter and artificial aerial cabinets are placed side by side and bolted together to form a single unit that encloses all parts carrying high RF or DC potentials.

Modulation of the carrier wave is carried out in the anode circuit of the RF power amplifier. The power modulator and its three stages of preliminary audio-frequency amplification are mounted in a single cabinet.

The first two audio-frequency stages are balanced class-A voltage amplifiers equipped with two tubes type EL 83 and QB 3.5/750, respectively. These are followed by the driver stage, containing eight tubes, type QB 3.5/750, connected in push-pull-parallel. This stage acts as a class A-B cathode follower and supplies the necessary driving power to the class B power modulator tubes, which are of the same type as those used in the final RF stage.

The cabinet in which the preliminary RF stages are housed, the RF power amplifier block and the modulator cabinet are lined up behind a common front as shown in Fig. 3.

Philips SOZ 294 fig. 4 As shown on the ground plan of Fig. 2, the modulating transformer, the modulating choke and blocking condenser and the anode smoothing filter are placed in a separate high-tension cubicle behind the transmitter. The space to the left of the transmitter is occupied by the anode voltage rectifier equipment. Low-tension distribution equipment and a set of three rectifiers supplying various voltages to the preliminary RF and AF stages are mounted on a common rack which is also placed behind the transmitter front.

Final RF stage

The final RF stage comprises three main elements, viz. the grid circuit and its coupling to the driver stage, the arrangement of the power amplifier tubes, and the anode circuit with its coupling to the harmonic suppression filter.

1 Grid circuit
As in its predecessor, the water-cooled transmitter, the grid circuit of the final amplifier has been designed as a Lecher system which, in order to save space, has been given the circular form shown in Fig. 4. This Lecher system is coupled to the driver stage by means of a parallel loop L1, as indicated on the circuit diagram of Fig. 5. In order to attain satisfactory tuning of the grid circuit over the entire frequency range, the latter has been subdivided into three sub-ranges. In the middle part of the range, running approximately from 9 Mc/s to 17 Mc/s, the shorting bridge between the two Lecher conductors includes a number of parallel-connected mica capacitors C1, whose impedance is negligible for this range. The presence of these capacitors permits separate measurement of the grid currents of the two tubes. For the lower part of the frequency range, from 5.9 to 9 Mc/s, the electrical length of the Lecher system becomes too small and a fixed vacuum capacitor C3 is connected directly in parallel to the grids of the tubes.

Philips SOZ 294 fig. 5 As the frequency increases above 17 Mc/s, the shorting bridge must be brought closer and closer to the grid connections of the tubes. Owing to the increase in frequency and the consequent increase in the current through capacitor C1 the losses in the shorting contacts go up and cause considerable heating. Moreover, the input capacitance of the tubes makes it impossible to attain resonance at the highest frequencies, while satisfactory coupling to the driver stage is also unattainable when the length of the Lecher system becomes too small.

For these reasons capacitor C1 is disconnected by means of the small servomotor, visible in Fig. 4, for frequencies above 17 Mc/s. In the range from 17 to 26.1 Mc/s the shorting bridge is constituted by vacuum capacitor C2, which has a relatively small capacity. This results in an electrical shortening of the Lecher system.

For the upper range of frequencies, from 24 to 26 Mc/s the shorting bridge again approaches the grid connections of the tubes. In this range the grid circuit may actually be considered as consisting of two Lecher systems which are coupled via capacitor C2, viz. the part to the right of C2 in Fig. 5 which is terminated by the tube capacitance, and the open Lecher system to the left of C2. Electrically, the length of this latter part is always smaller than a quarter wavelength, and its impedance, seen from the shorting bridge, always remains capacitive. As a result, the current through C2 only rises with frequency up to a certain point, after which it starts to decrease again. For the condition, not reached in this transmitter, where the open Lecher system would be exactly a quarter wavelength long the current through C2 would be reduced to zero. Proper dimensioning has made it possible to use this effect for obtaining low losses in the contacts and very smooth tuning at the higher frequencies, which is not critically affected by the position of the shorting bridge. The latter effect will be understood when it is noted that the increase in resonance frequency which results from moving the shorting bridge in the direction of the grids, is partially neutralized by the increase in capacitance of the open Lecher system.

Philips SOZ 294 fig. 6 2 Tube arrangement and neutralizing circuits
As shown by the diagram of Fig. 6, the two final amplifier tubes are placed symmetrically in their compartment, each anode cooler A standing on a ceramic support K, through which the cooling air enters and leaves the cooler. Fig. 7 is a close-up view of the upper part of the tube compartment.

It has been stated above that the final amplifier is of the grounded-cathode type.

In spite of the problems, to which we shall come back presently, connected with neutralizing an amplifier of this variety, problems which are largely absent in grounded-grid amplifiers, the former type has been chosen since it has a number of advantages:
  1. the driving power required is small;
  2. modulation can be restricted to the final amplifier, whilst with a grounded-grid amplifier extra modulation in the preliminary stages is required; this again gives rise to problems of phase shift which are avoided in the grounded-cathode amplifier;
  3. it is easier to attain a high efficiency of the final amplifier, and so to attain a good over-all efficiency of the transmitter.
For a push-pull amplifier neutralization is obtained in a simple manner by cross-wise capacitive coupling of the grids and anodes of the two tubes. If the neutralizing capacity is made equal to the anode-to-grid capacity of the tubes, the amplifier is theoretically stable for any frequency. In reality, however, the inductances of the connecting leads, and the stray capacities of the circuit elements make the adjustment of the neutralizing capacitor frequency-dependent.

For a push-pull amplifier neutralization is obtained in a simple manner by cross-wise capacitive coupling of the grids and anodes of the two tubes. If the neutralizing capacity is made equal to the anode-to-grid capacity of the tubes, the amplifier is theoretically stable for any frequency. In reality, however, the inductances of the connecting leads, and the stray capacities of the circuit elements make the adjustment of the neutralizing capacitor frequency-dependent.

Philips SOZ 294 fig. 7 In the present transmitter it has been found possible to use a single setting of the neutralizing capacitor over the entire frequency range, a setting which is not critically affected by tube replacements. This has been attained by giving the leads connecting the grid circuit to the grid terminals of the tubes and to the neutralizing capacitors a very definite inductance. The following considerations may make this clear:

Philips SOZ 294 fig. 8 Theoretically, the neutralizing bridge may be represented by the diagram of Fig. 8, in which, among other things, it has been assumed that:
  1. dimensions of elements are small compared to the wavelength, permitting the assumption of lumped inductances and capacities;
  2. the bridge contains only passive elements; the electron emission of the tubes is left out of consideration; this also applies to special effects such as the influence of space charges on the tube capacities;
  3. the mutual coupling between grid and filament can be neglected for the frequency range under consideration;
  4. corresponding elements in the various arms of the bridge, including the amplifier tubes, are nearly enough equal to permit considering the bridge as a symmetrical arrangement.
In the condition of equilibrium which we wish to attain the application of a HF potential between the anodes A1 and A2 of the tubes must not create a potential difference between the terminals B1 and B2 of the grid circuit and, for reasons of symmetry, these terminals must remain at earth potential. Neither must a potential be created between the grids g and the filaments ƒ of the tubes. In this condition of equilibrium the grid-to-filament capacitance Cgf may be considered as non-existent and the elements of the diagonal A1-A2 have no influence on the equilibrium of the bridge.

If B1 remains at earth potential, the condition that g1 and ƒ1 must show no potential difference leads to the equation: Cag.Lg = Caf.Lf.

In setting up the equations for the condition that B1 and B2 remain at earth potential, account must be taken of the considerable stray capacity Cv between the grid side of the neutralizing capacitor and earth. This can be done by carrying out a star-delta transformation which leads to a diagram of the shape indicated in Fig. 9. If B1 and B2 remain at earth potential, impedances ZAE and ZEB again have no influence on bridge equilibrium. ZBA is found to be: ZBA = jωLn (1 + Cv/Cn) + 1/jωCn, and is, therefore, equivalent to the series connection of the neutralizing capacitance Cn and an inductance equal to Ln (1 + Cv/Cn). It will be seen that B1 and B2 can only remain at earth potential, independent of frequency, when the following conditions are fulfilled:
Cag = Cn,
Lg = Ln (1 + Cv/Cn).
Philips SOZ 294 fig. 9 Combining equations (1) and (2), elements Lg, Ln and Cn can be evaluated. The value of the inductance Ln of the lead from the grid circuit to the outer electrode of the neutralizing capacitor is found as Ln = Lf Caf/(Cag + Cv).

If the stray capacity Cp is assumed to be approximately equal to capacity Cag, and the ratio Caf/Cag to have a customary value of 1/25, it is then found that the value of the inductance Ln must be approximately 50 times as small as the inductance Lf of the filament leads.

It has only been possible to attain such an extremely low value for Ln by means of an artifice. The connection between the grid circuit and the outer electrode of the neutralizing capacitor has been made to consist of a plane grid of parallel rods, the two grids being interwoven in such a manner that the magnetic coupling between the two grids results in and apparent inductance of the required low value. This value is set in the factory by adjusting the distance between the two outside rods of each grid and by varying the total number of rods. This design also makes it possible to carry currents of 100 to 150 A to the neutralizing capacitor

3 Anode circuit
In designing the anode circuit it appeared desirable to reduce the number of tuning elements to a minimum. In addition, the use of automatic frequency selection made it necessary to mount the Instantuners driving the relatively heavy tuning elements on the outside of the amplifier cabinet. For these reasons the anode circuit, like the grid circuit, has been designed in the form of a circular Lecher system, with the shorting bridge mounted on a rotating arm.

Philips SOZ 294 fig. 10 However, the much heavier currents circulating in the anode circuit have made it impossible to use shortening capacitors at the higher frequencies. Under these circumstances the high output capacitance of the transmitter tubes necessitated the choice of a very low characteristic impedance for the anode Lecher system in order to attain resonance at the highest radio frequencies. As a result the lowest frequency to which the Lecher system can be tuned without the parallel connection of capacitors is still comparatively high, viz 12 Mc/s. For lower frequencies capacitors C4 and C5 (see Fig. 5) are connected between the anodes. This is done in two steps which cover the frequency ranges from 12 to 8.5 Mc/s and from 8.5 to 5.9 Mc/s. In view of the low characteristic impedance of the Lecher system fairly high capacities had to be chosen for C4 and C5 and, as a result, the loaded Q of the anode circuit has become comparatively high for a broadcast transmitter of this power; depending on the frequency, it varies approximately between 22 and 45. Such a high loaded Q is an advantage, of course, in that it provides very effective suppression of the radio frequency harmonics generated by the power amplifier, but on the other hand the currents circulating in the anode circuit, which also pass via the shorting contact, become very high. In order to keep the anode circuit losses low the unloaded must therefore also be chosen very high. By a suitable choice of the dimensions for the anode Lecher system, shown in Fig. 10, circuit losses have been kept at a low figure. Depending on the frequency they reach a maximum of 1.5 to 2 kW, which is less than 2% of the carrier power. This volume of heat can easily be carried off by the natural air circulation in the amplifier cabinet.

In the water-cooled transmitter the cooling water was carried to the anodes of the transmitting tubes through the hollow conductors constituting the Lecher system and thereby provided excellent cooling of the contacts of the shorting stub. Naturally, this facility could not be maintained in the present transmitter and special care had therefore to be given to the design of the shorting contacts which, at certain frequencies, carry an effective current of approx. 600 A at 100% modulation.

As shown in Fig. 10, the shorting stub consists of a hollow box fitted on either side with a number of contact springs lying in the vertical plane. Each of these contact springs carries a contact of a special carbon-silver alloy. These two sets of contacts slide over the plane sides of the Lecher conductors which face each other. Cooling is obtained by means of a small flow of air of about 1 cu.m./min., conveyed through the hollow insulating arm which carries the shorting stub. This cooling is sufficiently effective to permit adjustment of the anode tuning while the transmitter is working at full power and 100% modulation. Coupling between the anode Lecher system and the aerial feeder matching network or the harmonic suppressor is by means of a hairpin-shaped coupling loop (L2 in Fig. 5), rotating in a vertical plane and driven by an Instantuner. In this manner a very simple design, requiring a minium of component parts, has been obtained.

Harmonic Suppression filter

According to the recommendations laid down by the Atlantic City Conference of 1947, the power of each harmonic supplied to the aerial feeder line by a broadcast transmitter should not exeed 200 mW. Owing to the high loaded Q of the final stage anode circuit this requirement is normally met by connecting a simple type of matching circuit between the transmitter output and the feeder line. However, for the transmitter illustrated in Fig. 3 the special requirement was laid down by the Netherlands PTT that the power of none of the harmonics should exeed 25 mW, which made the design of a special harmonic suppression filter necessary. This filter also includes a section for the effective suppresion of all harmonics falling within the first band of television frequencies (41-68 Mc/s). In addition to suppressing harmonics, the filter also serves to match the transmitter output circuit to the 320 ohm balanced impedance of the aerial feeder line. Matching remains possible for standing wave ratios of up to 1 : 1.5.

Philips SOZ 294 fig. 11 As shown on the diagram of Fig. 11, the harmonic filter proper consists of a single π-section comprising capacitors C1 and C2, and inductances L1. C1 and C2 are continuously variable vacuum capacitors, driven by a Multiturn Instantuner. The frequency range from 5.9 to 26.1 Mc/s has been divided into four sub-ranges and four sets of coils L 1 have consequently been provided. These are mounted on a turntable rotating on a vertical axis, which is driven by a servomotor and mounted in a separately screened section of the cabinet. Capacitors C1 and C2 are also mounted in separately screened compartments, of which that for C1 also houses the television interference filter. The latter filter comprises four series-resonant circuits composed of capacitors C3 and C4 and inductances L4 and L5, coupled via inductances L3. Inductances L2 serve to couple the television interference filter to the main harmonic suppression filter. The series-resonant circuits are tuned to two frequencies in television band I, for which the interference filter creates a domain of high attenuation.

Philips SOZ 294 fig. 12 For protection of the harmonic filter against excessive voltages resulting from feeder mismatch, a reflectometer unit has been fitted in the filter cabinet. As the diagram of Fig. 12 shows the reflectometer circuit is coupled capacitively and inductively to the feeder conductors by means of two loops L1 and L2. The inductive voltage vector is 180° out of phase with the capacitive voltage vector.

For a standing wave ratio of 1 : 1 on the feeder these two vectors cancel each other and no current flows through relay Re1. For higher standing wave ratios the two vectors no longer cancel, and current flows through Re1. This relay operates when the standing wave ratio reaches a figure of 1 : 2 and then trips the transmitter anode voltage.

Modulator

Full modulation of the carrier wave is attained with a modulator comprising four push-pull stages, viz. two voltage amplifiers, a driver stage that is connected as a cathode follower and a power stage. In spite of the large amount of negative feedback, approx. 23 db, from the anode circuit of the power amplifer to the grids of the tubes in the first amplifier, this limited number af stages provides a large degree of amplification. For a full 70 kW output from the modulator an input signal of approx. 300 mV suffices. In order that the negative feedback shall not give rise to instability, the first voltage amplifier, which is equipped with two type EL 83 tubes, contains a phase-correcting network. This stage also contains a simple clipper circuit, consisting of a rectifier bridge. Polarization of the rectifier cells is such that voltage peaks are cut off when the degree of modulation reaches a value of approx. 110%.

For 100% modulation the second voltage amplifier which is equipped with two type QB 3.5/750 tubes, produces a peak voltage of approx. 1000 V for driving the third amplifier stage. As is now customary for driver stages of this sort, the third stage is connected as a cathode follower. In order to yield sufficient current to the final modulator grids, the driver employs a number of QB 3.5/750 tubes in parallel. The advantages of cathode followers used as drivers for a power modulator have been described in an earlier article.

Two tubes of the same type as used in the final RF amplifier are employed in the power modulator. In order to obtain a low distortion figure the grid bias of each of these tubes can be adjusted separately.

For audio frequencies in the upper part of the range the impedance presented to the modulator is no longer real, but becomes complex, owing chiefly to the leakage reactance of the modulating transformer. Without counter-measures this would result in increased distortion and increased anode dissipation of the modulator tubes. For these reasons the anode impedance is corrected by means of a filter, inserted in the lead connecting the modulation transformer to the anode circuit of the RF power amplifier. This filter, which consists of a simple m-derived section, causes the phase angle of the anode impedance to increase less rapidly at the higher frequencies, and even to decrease beyond a certain frequency, while the factor of increase of the modulus remains below 1.5. As a result, the anode dissipation at 10.000 c/s is hardly larger than at 1000 c/s and a ventilator of relatively low power suffices to cool the modulator anodes. The low-pass character of the filter also assists in limiting harmonic distortion. Over the range from 30 to 10.000 c/s the linear distortion remains within ± 1 db, referred to the 1000 c/s point. Harmonic distortion at 100% modulation is approx. 0.5% from 100 to 2000 c/s. Near the boundaries of the pass-band the distortion rises to approx. 2%. Owing to the large amount of negative feedback intermodulation distortion is small, and the hum figure is better than - 65 db as compared to the 100% modulation level.

At the input of the modulator a low-pass filter with a cut-off frequency of 10.000 c/s and an attenuation of 40 db at 15 kc/s provides the necessary limitation of the transmitted bandwidth.

Power supplies

Power consumption and space requirements have been reduced to a minimum by limiting the number of rectifiers supplying power to the various RF and AF stages to four. This number includes the 10 kV anode rectifier for the RF and AF power amplifiers.

Power for the preliminary stages is supplied by a set of three rectifiers, indicated on the diagram of Fig. 1, which are mounted in a cabinet, together with the 380 V low-voltage distribution system. Grid voltages for all stages are derived from a 650V selenium rectifier, while another 800V selenium rectifier supplies the screen grid voltage for all preliminary stages and the anode voltage for all but the driver and final stages. The anode voltage for the driver stage tubes is supplied by a 4 kV rectifier employing three mercury vapour tubes, type DCG 6/18.

As stated, the rectifier cabinet also houses the low-voltage distribution system, which includes all low-voltage interlocking circuits, automatic starters for the filaments of the power tubes and some other control circuits. As shown on the layout of Fig. 2, this cabinet is placed in a conveniently accessible position behind the transmitter front.

By treating the 10 kV anode rectifier as an entirely separate unit, it has been possible to supply either a mercury tank rectifier or a rectifier employing mercury vapour tubes and thus meet all customer requirements in this respect. Normally, the transmitter is supplied with a rectifier employing six grid-controlled mercury vapour tubes, type DCG 7/100.

As usual for a rectifier of this type, voltage regulation from 0 to 10 kV is by electronic means and can be carried out by hand or fully automatically. Included in this regulating equipment is the high-speed protection against back-firing of the rectifer tubes and against short circuits and overloads originating in the transmitter proper. When an overload of this type occurs, the anode voltage is reduced to zero within 10 milliseconds and is then automatically brought back to its original value within a few seconds. If the overload persists, this cycle is repeated twice before the power circuit breaker definitely disconnects the rectifier.

Rectifier tubes and control equipment are mounted in a common frame, while the power transformer and smoothing filter are placed in a separate cubicle. The primary supply to the transformer is drawn directly from the high voltage mains: 10 kV, 50 c/s in the case under consideration.

Cooling system

As already mentioned, the tubes of the penultimate RF amplifier and those of the final RF and AF amplifiers have forced-air cooling. For the four type QBL 5/3500 tubes of the penultimate RF amplifier, which have radial coolers, the cooling air is supplied by a ventilator which is mounted on the floor of the preliminary RF amplifier cabinet. The type TBL 12/100 tubes of the final RF and AF amplifiers are provided with special coolers af a type that has been described in an earlier article. With these coolers the air, after passing along the anodes, is not blown into the transmitter cabinet, but flows into a downward outlet duct which brings the heated air out of the cabinet. As a result, no special measures are necessary for ventilating the final amplifier cabinets. For each of these, two ventilators are provided that are combined into one unit, as shown in Fig. 6. Normally, only one ventilator of each of these twin units is running, the second one acting as a spare. Cooling is then sufficient for the average 60% modulation. For special tests, when the transmitter is run with continuous 100% modulation, the two fans of each pair operate in parallel. Fig. 6 also shows how the ventilator unit contains a flap that shuts off the outlet of the ventilator which is not in service.

This flap is automatically blown into the correct position by the outlet air flow of the ventilator itself. When both ventilators are running, the air flow keeps the flap in the mid-position. Switching from one ventilator to the other therefore consists merely in pressing the start button of the correct ventilator motor. Careful design of these ventilators has resulted in a very low noise level.

Philips SOZ 294 fig. 13 Operational details

For normal routine operation all controls for starting up the transmitter are concentrated on the control desk shown on Fig. 13, The condition of the various interlocking contacts of the transmitter is signalled to a series of supervisory lamps on the control desk which provides a check on each step of the switching procedure. Any fault in the interlocking system can therefore be located at once. Essential controls are duplicated on the transmitter front panels and permit individual control of the various transmitter stages for readjustment or retuning. A number of measuring instruments are mounted on a horizontal panel running across the top of the transmitter front. Of these, the more important instruments are duplicated on the control desk. In order to provide the attendant with a check on the modulation quality, a number of monitoring points are connected to his desk. He is also given a visual check by means of a cathode ray tube showing an image of the modulated carrier at the input to the aerial feeder.

As stated earlier, the transmitting frequency may be automatically set to any one of six values in the full range from 5.6 to 26.1 Mc/s. For this purpose the tuning elements of the transmitter are driven by a number of single-turn and Multiturn Instantuners. Selection of the required transmitting frequency is by means of a switch mounted on the final RF amplifier. When this switch has been set to a new position, the transmitter is automatically tuned to the new frequency within a few minutes. Owing to this facility a change of frequency can be carried out by personnel with only a limited technical training.

In some cases it may become necessary to readjust the setting of a tuning element while the transmitter is in operation and without altering the adjustment of the automatic Instantuner driving that element. This happens, for instance, when the impedance of the aerial feeder changes owing to icing up. Later, when weather conditions become normal again and the feeder impedance returns to its original value, the Instantuner must be able to find its original setting again. For this reason the Instantuners permit correction of the setting of a tuning element without alteration of the 'spot' setting of the Instantuner.

TECHNICAL SPECIFICATIONS
Frequency range 5.95 to 26.1 Mc/s.
Carrier power 100 kW over the entire range. With slight modification the transmitter carrier power can be raised to 120 kW.
Audio frequency input level for 100% modulation Less than 0.775 V across 600 Ω.
Linear distortion Within ± 1 db from 30 to 10,000 c/s at a modulation of 60%.
Permissible continuous modulation 60%. Peak modulation 100%. Short splashes of overmodulation by 6 db can be tolerated.
Harmonic distortion


 from 50 to 1000 c/s : < 2%;
from 1000 to 3000 c/s : < 2.5%;
from 3000 to 10,000 c/s : < 3.5%.
Hum and noise level ≤-60 db, unweighted, referred to 100% modulation.
Over-all efficieny, unmodulated Approx. 55% for 100 kW, and nearly 60% for 120 kW carrier output.
Intermodulation distortion Two signals in the band from 5 to 10 kc/s that differ 180 c/s in frequency and each modulate the transmitter 30%, produce a percentage of intermodulation better than 2% for intermodulation products of the 2nd and 3rd order.

TUBE COMPLEMENT
RF stages AF stages and modulator Rectifiers
Number Type Number Type Number Type
2 TBL12/100 2 TBL12/100 6 DCG7/100
4 QBL5/3500 10 QB3.5/750 3 DCG6/18
4 QQE06/40 2 EL83

4 EL83





THIS TYPE OF TRANSMITTER IS INSTALLED IN THE FOLLOWING COUNTRIES

ITU Country
ITU Country
flag HOL HOL NETHERLANDS